The bidirectional DC-DC converter considered in this study is designed as a non-isolated battery-to-DC-bus power interface for electric vehicle charging applications. The converter employs a synchronous half-bridge configuration that enables bidirectional energy transfer between a lithium-ion battery and a higher-voltage DC bus. During the charging operation, the converter operates in buck mode to transfer energy from the DC bus to the battery, whereas during the discharging operation, it operates in boost mode to transfer energy from the battery to the DC bus.

The converter is designed for a rated power of 5 kW with a nominal battery voltage of 350 V and a DC-bus voltage of 450 V. A switching frequency of 5 kHz is selected for the semiconductor switches. The main design specifications are summarized in Table 1.

Table 1. Design specifications of the bidirectional DC-DC converter.

The rated battery current is determined from the rated converter power and nominal battery voltage as

During battery charging, the converter operates in buck mode. Under ideal steady-state conditions, the nominal buck duty ratio is expressed as

Therefore, the nominal duty ratio for buck operation is approximately 0.778.

During battery discharging, the converter operates in boost mode. The corresponding nominal duty ratio is

The nominal duty ratio for boost operation is approximately 0.222.

The converter is designed to operate in continuous conduction mode (CCM), where the inductor current remains continuous throughout each switching period. An allowable peak-to-peak inductor-current ripple of 20% of the rated battery current is selected:

For buck operation, the required inductance can be determined from

Hence, the calculated inductance is approximately 5.5 mH. The selected inductance limits the current ripple while supporting CCM operation at the rated operating point. The validity of the CCM assumption is subsequently verified from the simulated inductor-current waveform in the Results and Discussion section.

The DC-link capacitor is used to reduce the voltage ripple at the DC bus and to provide short-term energy buffering during converter switching. The capacitance is designed based on the allowable DC-bus voltage ripple during boost operation, which represents the battery-to-DC-bus power-transfer condition. The average DC-bus current at the rated power is calculated as

The allowable DC-bus voltage ripple is selected as 1% of the nominal DC-bus voltage:

For boost-mode operation, the minimum DC-link capacitance can be approximated by

A capacitance higher than the calculated minimum is selected to provide additional margin for DC-bus voltage-ripple suppression. Therefore, the selected DC-link capacitance is 220 μF. The selected value is approximately twice the calculated minimum capacitance and provides additional filtering margin during switching and bidirectional power-flow transition.

A capacitor is also connected across the battery-side terminal to attenuate high-frequency switching components and reduce the voltage ripple applied to the battery. Unlike the DC-link capacitor, the battery-side capacitor is sized based on the inductor-current ripple and the allowable battery-terminal voltage ripple. For the present design, the allowable battery-side voltage ripple is selected as 0.5% of the nominal battery voltage:

Assuming a triangular inductor-current ripple, the minimum battery-side capacitance can be estimated as

The required capacitance becomes 40.86 μF. A higher practical value is selected to provide sufficient filtering margin; we choose 100 μF. The selected battery-side capacitor reduces the high-frequency voltage ripple at the battery terminals without directly determining the average battery charging or discharging current. A small series resistance of 1 × 10⁻⁴ Ω was included in the passive branches to represent non-ideal component characteristics and facilitate numerical simulation. The calculated and selected converter parameters are summarized in Table 2.

Table 2. Calculated and selected design parameters of the bidirectional DC-DC converter.

Figure 1 shows the MATLAB/Simulink implementation of the proposed non-isolated bidirectional DC-DC converter based on the calculated design parameters. The model was implemented with a discrete simulation step of 1 μs.

Figure 1. MATLAB/Simulink model of the bidirectional DC-DC power converter.

Battery Model and Parameters

A lithium-ion battery model was employed to represent the EV battery pack in the simulation. The battery was configured with a nominal voltage of 350 V and a rated capacity of 100 Ah, corresponding to a nominal energy capacity of 52.5 kWh. The battery energy capacity is expressed as

where V b a t is the nominal battery voltage and Q b a t is the rated battery capacity. Thus, E b a t = ( 350 ) ( 100 ) = 35   kWh .

An initial SOC of 60% was selected to provide sufficient operating range for evaluating both charging and discharging conditions. The battery response time was set to 30 s, while temperature and aging effects were excluded to focus on the electrical behavior of the proposed converter. The battery parameters used in the simulation are summarized in Table 3.

Table 3. Battery model parameters.

The battery terminal voltage varies with the battery operating condition rather than remaining fixed at its nominal value. Consequently, the measured terminal voltage may differ from the nominal value of 350 V during charging and discharging due to the battery discharge characteristics, SOC, internal resistance, and the direction and magnitude of the battery current. The SOC was obtained directly from the built-in lithium-ion battery model in MATLAB/Simulink based on the battery current and specified battery capacity; no separate SOC estimation algorithm was implemented in this study.

The proposed bidirectional DC-DC converter operates in two modes according to the direction of power flow between the DC bus and the EV battery. Referring to the converter topology in Figure 1, charging is achieved through buck operation, whereas discharging is achieved through boost operation. The direction of operation is determined by the battery-current reference applied to the closed-loop current controller. In this study, the battery-current sign convention is defined as

The general operating principle of a non-isolated bidirectional DC-DC converter is illustrated in Figure 2. The schematic is intended to provide a simplified representation of the bidirectional power-flow mechanism rather than the exact configuration of the proposed simulation model shown in Figure 1. In general, charging is achieved through buck operation with power transferred from the DC bus to the battery, whereas discharging is achieved through boost operation with power transferred from the battery to the DC bus.

Figure 2. General operating principle of a bidirectional DC-DC converter: (a) charging (buck mode) and (b) discharging (boost mode).

2.2.1. Charging Operation (Buck Mode)

During charging, power is transferred from the 450 V DC bus to the EV battery. The converter operates in buck mode to reduce the DC-bus voltage to the required battery-side voltage while regulating the charging current according to the specified negative current reference. Energy transfer is controlled through the complementary switching action of the two semiconductor switches. During charging, power is transferred from the 450 V DC bus to the EV battery. Since the DC-bus voltage is higher than the battery voltage, the converter operates in buck mode. The two semiconductor switches are driven by complementary PWM signals to regulate the battery current according to the specified positive current reference. Based on the adopted sign convention, I b a t > 0 represents battery charging.

2.2.2. Discharging Operation (Boost Mode)

During discharging, the direction of power flow is reversed, with energy transferred from the EV battery to the higher-voltage DC bus. The converter therefore operates in boost mode, where the battery-side voltage is stepped up to the DC-bus voltage. Based on the adopted current sign convention, a negative battery current, I b a t < 0 , represents a discharging operation. The battery current is regulated according to the specified positive current reference through the closed-loop current controller. The complementary switching action of the semiconductor switches controls the energy stored and released by the inductor, thereby enabling regulated power transfer from the battery to the DC bus.

A closed-loop PI current controller is employed to regulate the battery current during both charging and discharging operations. The control structure uses the battery-current reference, I r e f , as the command input, while the measured battery current, I b a t , is fed back to the controller. The current error is defined as

The current error is processed by a continuous-time PI controller to generate the duty-cycle command. The controller transfer function is expressed as

where K p and K i are the proportional and integral gains, respectively, E ( s ) represents the current error in the Laplace domain, and D represents the duty-cycle command supplied to the PWM generator. The PI controller output, representing the duty-cycle command, was constrained within the range 0 ≤ D ≤ 1 prior to PWM generation, corresponding to a duty-cycle range of 0% - 100%.

The PI controller gains were determined using an iterative trial-and-error tuning approach. The proportional gain, K p , was initially adjusted to obtain a sufficiently fast current-tracking response, while the integral gain, K i , was subsequently tuned to minimize the steady-state current error. The gains were iteratively refined by observing the transient response, steady-state error, and current oscillations under both charging and discharging conditions. The same PI current controller is employed for both charging and discharging operations, with the direction of power flow determined by the sign of the reference current as defined in (12). The PI controller parameters used in this study are summarized in Table 4.

Table 4. PI current-controller parameters.

The duty-cycle command generated by the PI controller is subsequently converted into the PWM signal, PWM1, which is applied to the upper semiconductor switch, S1. The complementary switching signal, PWM2, is generated from PWM1 and applied to the lower switch, S2. This complementary switching arrangement ensures coordinated operation of the two semiconductor switches and enables controlled energy transfer through the converter inductor. The switching action is continuously adjusted according to the duty-cycle command produced by the PI controller in response to the battery-current error. Consequently, the measured battery current is regulated toward the specified reference current during both charging and discharging operations. The conceptual closed-loop current-control structure is illustrated in Figure 3.

Figure 3. Closed-loop PI current-control structure of the bidirectional DC-DC converter.

The corresponding MATLAB/Simulink implementation of the current-control scheme is shown in Figure 4.

Figure 4. MATLAB/simulink implementation of the proposed PI current controller.

Figure 4 presents the MATLAB/Simulink implementation of the proposed closed-loop current-control scheme. The current-reference selector is used to switch between the predefined charging and discharging current references. The selected reference, I r e f , is compared with the measured battery current, I b a t , and the resulting current error is processed by the PI controller. The controller output determines the duty-cycle command for the 5 kHz PWM generator, which produces PWM1 for the upper switch. The complementary signal, PWM2, is generated from PWM1 and applied to the lower switch. This implementation enables a single PI current-control loop to regulate the battery current during both directions of power transfer.

During mode reversal, the current-reference selector changes the battery-current reference between the charging and discharging commands while the same PI current controller remains active. For the rated operating condition considered in this study, the current reference is constrained within −14.3 A to +14.3 A. No explicit integrator reset or hold mechanism is applied during the transition. The complementary switching signal, PWM2, is generated directly from PWM1.

An explicit dead-time circuit is not included in the present simulation model. Therefore, the present switching implementation is intended to evaluate the converter-level current-control behavior, while practical dead-time implementation and associated switching-device protection are beyond the scope of this simulation study.

Figure 5 presents the converter response for a current-reference reversal from −5 A to +5 A. During the first 5 s, the measured battery current remains close to −5 A, demonstrating stable current regulation during discharging. At t = 5 s, the reference is changed to +5 A, and the battery current rapidly follows the new operating command and remains close to the desired charging current. No sustained oscillation is observed following the mode reversal.

The battery terminal voltage responds consistently to the change in current direction. During the discharging interval, the terminal voltage decreases gradually from approximately 377.6 V to 377.4 V. Following the transition to charging at t = 5 s, an increase in terminal voltage is observed, after which the voltage gradually rises during the charging interval. The SOC response provides further confirmation of the bidirectional power-flow behavior. The SOC decreases during discharging and reaches its minimum value at the mode-reversal instant before increasing during charging. The relatively small SOC variation is expected because of the 100 Ah battery capacity and the short simulation interval.

Figure 6 shows the corresponding responses for a current-reference reversal from −10 A to +10 A. The measured battery current closely follows the commanded value in both operating modes. Prior to t = 5 s, the current is maintained at approximately −10 A during discharging. Following the reference reversal, the current changes to approximately +10 A and remains regulated around the charging-current reference. The response indicates that the same PI controller can regulate the battery current at a higher current magnitude without requiring controller retuning.

Figure 5. Battery responses during current-reference reversal from −5 A to +5 A: (a) battery current, (b) battery terminal voltage, and (c) SOC (%).

Figure 6. Battery responses during current-reference reversal from −10 A to +10 A: (a) battery current, (b) battery terminal voltage, and (c) SOC (%).

Compared with the ±5 A case, the higher current magnitude produces a more noticeable change in battery terminal voltage and SOC. The terminal voltage decreases during the discharging interval and exhibits an upward change when the converter switches to charging mode. Similarly, the SOC decreases more rapidly during discharging and subsequently increases during charging. These trends are consistent with the increased magnitude of battery current and, therefore, the higher rate of energy transfer.

Figure 7 presents the converter performance at the rated current-reference magnitude of 14.3 A. During the initial discharging interval, the battery current is regulated close to −14.3 A. At t = 5 s, the reference is reversed to +14.3 A, producing a rapid transition from discharging to charging operation. A small transient is observed immediately after the mode reversal before the current converges to the commanded charging level. The current subsequently remains stable around the reference value, demonstrating that the PI controller is capable of regulating the converter at its rated current condition.

Figure 7. Battery responses during current-reference reversal from −14.3 A to +14.3 A: (a) battery current, (b) battery terminal voltage, and (c) SOC (%).

The effect of the higher current magnitude is also evident in the battery terminal voltage. During discharging, the voltage gradually decreases and reaches its lowest value immediately before the mode transition. When the current direction is reversed, the terminal voltage increases abruptly due to the change in current direction and subsequently rises gradually during charging. The SOC exhibits the same expected trend, decreasing from its initial value during discharging and increasing after the transition to charging. Among the three investigated operating conditions, the ±14.3 A case produces the largest SOC variation because it corresponds to the highest battery-current magnitude.

Overall, the results in Figures 5-7 demonstrate consistent bidirectional current regulation over the investigated range of ±5 A to ±14.3 A. The same PI controller successfully tracks the different current-reference levels and maintains stable operation before and after the discharging-to-charging transition. Increasing the current-reference magnitude results in greater changes in battery terminal voltage and SOC, as expected due to increased power transfer. More importantly, the measured battery current matches the commanded current direction and magnitude across all three operating conditions, demonstrating the effectiveness of the proposed closed-loop current-control strategy for the DC-bus battery interface investigated.

The switching behaviour and inductor-current characteristics were examined under the rated current condition of 14.3 A for both power-flow directions. Figure 8 and Figure 9 present the PWM signals applied to the upper and lower semiconductor switches together with the corresponding inductor current, I L , during discharging and charging operations, respectively. The two gate signals exhibit complementary switching behaviour, whereby PWM2 is the logical complement of PWM1. This switching pattern is consistent with the PWM implementation described in Section 2.3.

Figure 8 shows the switching characteristics during rated discharging operation at I r e f = − 14.3   A . The inductor current remains negative and fluctuates around the rated discharging-current level. Although switching ripple is present, I L remains continuous throughout the observed interval and does not approach or cross zero. This indicates that the converter operates in continuous conduction mode (CCM) under the rated discharging condition.

Figure 8. Switching and inductor-current characteristics during rated discharging operation at I_ref = −14.3 A: (a) PWM1, (b) PWM2, and (c) inductor current I_L.

Figure 9. Switching and inductor-current characteristics during rated charging operation at I_ref = +14.3 A: (a) PWM1, (b) PWM2, and (c) inductor current I_L.

Figure 9 presents the corresponding behaviour during rated charging operation at I r e f = + 14.3   A . The polarity of the inductor current is reversed compared with the discharging condition, with I L remaining positive around the rated charging-current level. Similar to the discharging case, the inductor current remains continuous throughout the switching interval without reaching zero. Therefore, CCM operation is also maintained during rated charging.

The results in Figure 8 and Figure 9 demonstrate that continuous inductor-current operation is maintained in both directions of power flow at the rated current condition. The current polarity changes according to the operating mode, while the complementary switching strategy remains unchanged. These results support the suitability of the selected converter parameters for bidirectional operation under the investigated rated condition.