Study and Enhanced Design of RF Dual Band Bandpass Filter Validation and Confirmation of Experimental Measurements

doi:10.4236/cs.2011.24041

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Circuits and Systems, 2011, 2, 293-296

doi:10.4236/cs.2011.24041 Published Online October 2011 (http://www.scirp.org/journal/cs)

Study and Enhanced Design of RF Dual Band Bandpass

Filter Validation and Confirmation of Experimental

Measurements

Mohamed Mabrouk, Leila Bousbia

CIRTACOM and ISETCOM of Tunis, University of Carthage, Tunis, Tunisia

E-mail: [email protected]

Received July 22, 201 1; revised August 12, 2011; accepted August 21, 2011

Abstract

Dual band bandpass filter is designed and optimized for RF wireless applications. The performances of that

RF dual band filter are improved especially parameters describing the insertion loss, return losses and rejec-

tion. Dual band bandpass filter using stub loaded resonators is designed and characterized. Theoretical re-

sults are compared with experimental data. This comparison shows that the magnitude of reflection coeffi-

cient S11 from ADSTM simulation is better than 28.0 dB, and the insertion loss S21 is less than 0.5 dB. The

two rejections are also better than 32.0 dB. The simulated results also show that two central frequencies are

located at desired 1.82 and 2.95 GHz. Comparison of measured and simulated results shows frequency drift.

The main reason for this frequency shifting is due to some uncertainties. These are obviously due to geomet-

rical and physical parameters H and r respectively of Duroid substrate used during design and measure-

ments.

Keywords: RF Filters, Dual Bandpass, Effective Permittivity, Physical and Geometrical Parameters

1. Introduction

RF circuits with a dual pass band operation are required

by modern wireless communication systems [1]. For

example, the RF transceiver length (TX/RX) for second

generation GSM and third-generation WCDMA mobile

communications must be able to receive and transmit

900 MHz and 1900 MHz signals. Also 2.4 GHz and 5.2

GHz bands are two bands operated by high-speed

wireless LANs [1,2]. The band pass filter is necessary to

generate two or more transmission frequency bands, in

particular dual-band filters, as they hav e an essential role

in transmit-receive systems [2]. RF filters have par-

ticularly an important role for signals filtering, rejection

and isolation between parts in transceivers systems

(TX/RX). Filters have the characteristic to be frequency

selective devices for transmitting and attenuating signal

in desired frequency ranges [3]

Many different ways are considered to design dual-

band bandpass filters such combination of two filters

operating on two different bandwidths [4] and using dual

band stepped impedances resonator [5] or dual band stubs

resonators [6]. Quasi lumped with open loop band pass

filters operating at different frequency bands [7] and

using square loop dual mode resonators [8] are also used

for designing this type of filters. Small insertion loss, low

return loss and high rejection band are the desired

characteristics of a good filter. The design of this kind of

filter is considered using electromagnetic (EM) simu-

lators. The studied filter was simulated usin g IE3DTM [6],

and in this paper we report our ADSTM simulation

optimized results which are very close to the experimental

measurements. ADSTM software has been used succe-

ssfully many times for simulation design filter so th at the

resulting performances meet the microwave filter speci-

fications. Simulated results confirm that two central

frequencies are located at desired values 1.82 GHz and

2.95 GHz.

2. Design Description of Band Pass Filter

The studied filter is composed of two ring-resonators

loaded with two open stubs. Figure 1 shows the physical

layout of the dual band pass filter using uniform

microstrip lines. This filter consists of two microstrip

open loops. Two open circuited stubs are attached [6] at

M. MABROUK ET AL.

294

the center of the respective microstrip lines. The total

length of each resonator is around half wave length

(



g/2).

The design parameters dimensions are chosen [6] as

following: W = 1.2 mm; W1 = 1.5 mm; W2 = 0.9 mm;

L=11.2 mm; L1 = 6.7 mm; g1 = g2 = 0.3 mm. The struc-

ture of open-loop resonators filter with center frequency

of 1.82 GHz and 2.94 GHz is de signed on an RT Duroid

6006 substrate with a thickness H = 0.635 mm ± 0.0254

and a relative dielectric constant



r = 6.15 ± 0.15 [9].

3. Analysis and Comparison of Simulation

with Measurement Results

Figures 2 and 3 illustrate simulated and measured S21

and S11 parameters respectively of our studied dual band

filter.

Simulated results show that two central frequencies

are located at the desired values 1.82 GHz and 2.95 GHz,

and a small frequency shifting is observed. Thus, the

ADSTM simulation results are shifted of –5.0 MHz,

(1.825 GHz instead 1.830 GHz), and of +13.0 MHz

(2.953 GHz instead 2.940 GHz). To understand the main

reason for this frequency shifted aberration which is

certainly due to the uncertainties on the guided wave

length



g, we have highlighted the following rela-

tionships:

eff



ε 1

where F is the center frequency of the filter and c is the

light velocity. The effective dielectric constant is de-

pending on relative permittivity



r and ratio WH be-

tween the width W of the transmission line and the

thickness H of the substrate which is given in the fol-

lowing forms. The RT Duroid 6006 laminate substrate

is available with relative permittivity



r value



r = 6.15

± 0.15.

It’s known that the Equation (2) shows below the

effective permittivity



eff of the microstrip structure, used

Figure 1. Layout of dual band filter.

0.5 11.5 22.5 33.5 44.5

-70

-60

-50

-40

-30

-20

-10

Simulation

Measurement

Figure 2. Simulated and measured S21 parameters of dual

band filter.

0.5 11.5 22.5 33.5 44.5

-50

-45

-40

-35

-30

-25

-20

-15

-10

-5

Simulation

Mea surem ent

Figure 3. Simulated and measured S11 parameters of dual

band filter.

in our design, depends on the relative dielectric constant



r, which is one of physical parameters of Duroid

substrate, and the guided wave length



g also depends on



eff, so the resonant frequencies of our filter are closely

depending on g according to the Equation (1).

For WH  1, the empirical relationship of the

effective permittivity is summarized below [10-12]:

eff 11 H

112

22 W









 





ε



2

Because of the uncertainty on the dielectric relative

permittivity



r of Duroid substrate recalled previously

(in §2), we can evaluate



eff as following :

eff

4.476 4.553





 3

M. MABROUK ET AL.295

76 mm78 mm





4

Thus, we can deduce that the first resonant fre-

quency FC1 is the following

1.79 GHzF1.84 GHz 5

For the second frequency FC2, we can also do the

same line of argument as following

47mm48 mm  6

That means FC2 is:

2.94 GHzF3.06 GHz 7

These equations confirm the first explicit reason for

this frequency shifting is due to the uncertainty on the

first physical parameter, i.e. the dielectric relative per-

mittivity



r of Duroid substrate.

In the same line of argument, we can see that the

uncertainty on the substrate thickness H can lead the

following limits for the first frequency FC1:

eff

4.47 4.507



 8

77mm78mm





9

1.810 GHzF1.850 GHz 10

For the second frequency FC2:

eff

4.47 4.507



 11

47mm48 mm





12

2.910 GHzF3.0 GHz 13

Here again, it’s confirmed that the uncertainty on

thickness H can obviously contribute to the frequency

shifting that we observed, and the differences between

center frequencies are surely due to the cumulative

effects of both uncertainties on



r and H. We have

made a comparative study of our simulation results and

experimental measurements obtained and provided by

our partner Zhang. Nevertheless, these differences that

we observed on the frequency values, between our

simulations and the measurements of our partner

Zhang, are remaining very small. Thus, the difference

of 5.0 MHz on the first resonance frequency (1.830

GHz) between our simulation results and the mea-

surements of Zhang is about 0.27%, while the dif-

ference of 150.0 MHz observed (1.680 GHz instead

1.830 GHz) between the first simulation results of

Zhang and the experimental measurements is about

8.0%. The difference of 13.0 MHz on the second

resonance frequency (2.94 GHz) between our simu-

lation results and the measurements of Zhang is about

0.44%, and the difference of 130.0 MHz observed by

Zhang (2.81 GHz instead 2.94 GHz) is about 4.4%.

The little disagreement we have obtained with the

experimental measurements confirms our predictions,

and our indepth analysis shows that we can notably

improve some characteristics of the studied filter with

going in detail and making further development, con-

sequently of the global performances become en-

hanced.

Regarding the insertion and return losses, Figure 4

shows simulated and measured insertion loss S21 and

return loss S11 parameters respectively of our dual band

filter. We obtained S21 significantly lower than 0.28 dB

instead of 0.9 dB measured at FC1, and 0.4 dB instead

of 1.1 dB measured at FC2 respectively. Our simulated

S21 results depicted on Figure 2 are up to 10.0 dB

better at 4.0 GHz than the measurements. Figure 3 also

shows that our simulated results of return loss S11 are

up to 8.0 dB better at 4.0 GHz than the measurements,

and the obtained return loss S11 is better than 28.0 dB.

Moreover, the simulated filter using two open loop

ring resonators shows better rejections than the meas-

ured ones, 31.0 dB instead of 28.0 dB measured at 0.5

GHz and 37.0 dB instead of 31.0 dB measured at 4.5

GHz. Table 1 shows comparative results.

4. Conclusions

We have studied and enhanced the design of dual band

bandpass filter for RF and wireless applications. The

performances of RF dual-band filter are improved

especially parameters describing the insertion loss, return

losses and rejections. We have obtained a good agree-

ment between our simulations and experimental results.

Insertion loss S21 lower than 0.5 dB and return loss S11

better than 28.0 dB were obtained from ADS simulations.

The rejections are also better than 32.0 dB. Simulated

results show that two central frequencies are located at

desired 1.820 and 2.950 GHz. Comparison of measured

and simulated results shows frequency shift. This is

obviously due to the uncertainties on the geometrical and

Table 1. Comparison of Zhang measurements [5] and our

simulations.

Measurement

Results

of Zhang [5] Our results

Resonance Frequency FC1 1. 830 GHz 1.825 GHz

Resonance frequency FC2 2.940 GHz 2.953 GHz

Band Pass 1 9.4% 6.7%

Band Pass 2 7.5% 4.37%

Insertion losses at FC1 0.9 dB 0.28 dB

Insertion losses2 at FC2 1.1 dB 0.4 dB

Return loss at FC1 24.0 dB 29.25 dB

Return loss at FC2 20.0 dB 28.81 dB

Rejection at 0.5 GHz 28.0 dB 31.64 dB

Rejection at 4.0 GHz 31.0 dB 38.64 dB

M. MABROUK ET AL.

296

0.5 11.5 22.5 33.5 44.5

-70

-60

-50

-40

-30

-20

-10

Frequency [GHz]

Magnitude [dB]

Simulation

Meas ure me nt

Figure 4. Simulated and measured S11, S21 parameters of dual-band filter.

physical parameters H and



r respectively of Duroid

substrate used during design and measurements.

5. Acknowledgements

The authors would like to acknowledge and extend their

gratitude to our partner Xiu Yin Zhang, Associate Pro-

fessor with the School of Electronic and Information

Engineering (South China University of Technology,

Guangzhou, 510641, China) who has carried out and

provided the experimental measurements of studied pro-

totype of filter.

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